Automatic Antenna Tuning Unit to Improve RFID System Performance

IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 60, NO. 8, AUGUST 2011 2797 Automatic Antenna Tuning Unit to Improve RFID System Performance Hannes Wegleiter, Bernhard Schweighofer, Christian Deinhammer, Gert Holler, Member, IEEE, and Paul Fulmek Abstract—Systems for radio-frequency identification (RFID) realize a bidirectional wireless communication between a reader device and cheap passive tags (e.g., an RFID chip with an antenna printed on badges for access-control systems) in the near field of an antenna. This paper deals with the automatic and fast tuning of the resonant reader antennas in the widely used 13.56-MHz frequency band, resulting in a performance improvement and an increased READ / WRITE range. A novel RFID reader system is proposed with a tuning transformer for the automatic adjustment of the resonant circuit center frequency. This is achieved by shifting the magnetic operating point (i.e., the biasing direct-current (dc) magnetization) of a Ni–Zn ferrite transformer by means of an additional dc winding. The operation principle and the performance potential of this device are demonstrated by measurements on a prototypical RFID system. Index Terms—Antenna accessories, impedance matching, loop antennas, magnetic devices, magnetic-variable control. I. I NTRODUCTION ADIO-FREQUENCY identification (RFID) [1] systems are based on wireless communication between the RFID tag and reader in the near field of an antenna. The bidirectional communication is realized by magnetic coupling between the reader and the tag (i.e., air-transformer principle). The tags consist of low-power microelectronic circuitry combined with a printed coil used as an antenna. The data on the tag are usually read by a stationary or handheld reader device consisting of a high sensitive receiving electronic, a powerful transmitting stage, and an antenna. The required power for the RFID tag is supplied through the electromagnetic field of the reader antenna. Depending on the frequency range used, different READ ranges can be achieved. RFID systems are commonly used in various applications such as fare payment in the public transportation sector, electronic ticketing, animal identification, access control, industrial measurement, control applications, and logistics. RFID systems in the fields of part and stock identification can help ease quality R Manuscript received June 18, 2010; revised August 23, 2010; accepted October 14, 2010. Date of publication May 2, 2011; date of current version July 13, 2011. This work was supported by the Austrian Science Fund (FWF) under Reference L356-N14. The Associate Editor coordinating the review process for this paper was Dr. Mark Yeary. H. Wegleiter, B. Schweighofer, C. Deinhammer, and G. Holler are with the Institute of Electrical Measurement and Measurement Signal Processing, Graz University of Technology, 8010 Graz, Austria. P. Fulmek is with the Institute of Sensor and Actuator Systems, Vienna University of Technology, 1040 Vienna, Austria. Color versions of one or more of the figures in this letter are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIM.2011.2122390 control tasks while increasing the flexibility of production processes. In many RFID applications, handheld devices are used as readers. Accordingly, the positions and the orientations of the antenna, the tag, and the objects in the environment may continuously change in a wide range, with respect to each other. The tuning of the whole electromagnetic system is influenced, resulting in changed resonant frequency, quality factor Q, and bandwidth [2]. Consequently, the efficiency and the READ range of the RFID system are influenced. In this paper, a novel reader based on an electrically tuned antenna transformer is presented, which is capable of compensating for the influences of changing objects close to the antenna, thus achieving a constant high reading performance. In the succeeding sections, the operating principle and the tuning transformer are described. The achievable tuning range is compared with the required tuning range for typical RFID systems with handheld readers. The measurement setup for system evaluation is presented, and the obtained measurement results are discussed. II. O PERATING P RINCIPLE Passive RFID tags are powered by the electromagnetic field in the near field of the reader antenna [3]. To provide the required magnetic-field strength, the antenna is driven by a power amplifier. Typically, the communication link between the reader and the tag is realized using the amplitude modulation of the carrier while the back transfer is performed by load modulation. The RFID tag responds by modulating the resistive load of its antenna circuit resulting in a minor change in the electromagnetic field of the reader antenna. This field modulation induces a voltage on the reader antenna and can be measured by means of a highly sensitive amplification circuit. For an optimal power transmission between the reader and the tag, a resonant circuit tuned to the carrier frequency of 13.56 MHz is realized. A block diagram of the realized RFID reader is shown in Fig. 1. The setup consists of a high-power transmitter stage combined with a sensitive demodulation unit and a series resonant circuit. The effective inductance at the input of the series resonant circuit is composed of the inductances of the loop antenna and the tuning transformer. Resistor R determines the quality factor Q of the resonant circuit. This quality factor directly influences the received signal strength. Therefore, it should be selected as high as possible. On the other hand, the higher the quality factor Q of the resonant circuit is chosen, the more the bandwidth of the resonant circuit is reduced, and the sidebands of the communication channel are suppressed. A good compromise is a value of Q ≈ 30. 0018-9456/$26.00 © 2011 IEEE 2798 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 60, NO. 8, AUGUST 2011 Fig. 1. Block diagram of the complete RFID setup. The heart of the RFID reader is the antenna tuning transformer. The tuning unit controls the dc magnetization of the ferrite antenna transformer (ring core) and thus adjusts the resonant frequency of the whole antenna circuit. The unit can compensate for environmental changes, component tolerances, and aging in a wide range. For RFID applications using a handheld reader device (e.g., animal identification, inventory systems, and industrial monitoring), the quality factor is usually reduced since the movement of the handheld device causes continuous variations of the parameters characterizing the communication link between the reader and the RFID tag. These variations lead to a shift of the resonant frequency of the reader. For systems with high Q factor and narrow bandwidth, this frequency shift results in a strongly dampened signal, due to the mismatch of the resonant frequencies of the reader and the tag, thus drastically reducing the reading range. Hence, conventional readers without an automatic antenna tuning use a lower Q factor to increase the bandwidth of the system. The lowered Q factor, however, implies a decreased gain of the resonant circuit and a reduced maximum reading range. To realize a robust antenna system with a high Q factor and a large reliable reading range for handheld devices, we present in this paper a resonant circuit with a continuous real-time tuning capability. Furthermore, tolerances of components (temperature effects, component aging, etc.) can be compensated by tuning the resonant circuit. The tuning transformer contained in the circuitry (see Fig. 1) fulfills the requirements concerning continuous tuning capability and maintenance-free operation, as well as compact and cost-efficient design. Therefore, the electrically tunable inductance (transductor; see also [4]) approach is to be preferred compared with switched and electromechanical devices. Fig. 2. Hysteresis loop of a NMG M2 ferrite [6]. Minor loops with ∆B = 10 mT in different operating points are shown. The straight extrapolated lines indicate the effective differential permeabilities of these lancets [5]. III. T UNABLE T RANSFORMER Fig. 3. Principle layout of the tuning transformer with primary and secondary RF windings on a ferrite toroid and the dc control winding on an iron core. The working principle of the tunable transformer is based on the nonlinear properties of the ferrimagnetic core material. Typical RF signals cause only minor excitations of the ferrite and can be described by the small-signal behavior of the material (i.e., incremental or differential permeability in the actual operating point). By applying a direct-current (dc) premagnetization to the ferrite material of the inductance, the operating point can be changed, and the differential permeability relevant for the small-signal behavior can be adjusted (compare Fig. 2 [5]). Thus, the inductance of the device (a so-called transductor) can be electrically adjusted. Based on the principle of an adjustable inductance, the component can be extended by adding a secondary winding. This leads to an adjustable transformer offering a flexible impedance transformation functionality. The device can be used for choosing the appropriate transformation ratio as required by the antenna setup for the reader. Moreover, the balancing of an unbalanced signal can be achieved by grounding the middle tap of the secondary winding. The basic layout of the tuning transformer with the primary and secondary RF windings on a ferrite toroid is shown in Fig. 3. In contrast to commonly used approaches (e.g., [7]), the control winding needed for the premagnetization is located on a separate iron core and not on the ferrite torus itself. Although this design requires more complex manufacturing steps, it efficiently decouples the dc premagnetization and the RF flux. Fig. 4 shows the results of a 2-D simulation of one quarter of the tuning transformer. The magnetic flux lines for WEGLEITER et al.: AUTOMATIC ANTENNA TUNING UNIT TO IMPROVE RFID SYSTEM PERFORMANCE 2799 Fig. 6. Construction of the tuning transformer consisting of a dc coil placed on a steel yoke and the RF coils placed on the ferrite torus. Dimensions are in millimeters. Fig. 4. Magnetic flux lines in the tuning transformer with dc excitation in the control winding. The whole flux generated by the dc in the control coil is carried by the ferrite core [4]. the material. Due to the very small skin depth of only a few micrometers in the iron yoke, the losses due to the eddy currents can be neglected. For the finite-element method simulation in Figs. 4 and 5, the relative permeability was chosen as 125 for both materials [8]. The specific electrical resistivity of the ferrite core is 105 Ω · m; the resistivity of the steel yoke is orders of magnitudes lower, i.e., 10−6 Ω · m. The actual design of the tuning transformer used for the simulation and the experiments is shown in Fig. 6. The ferrite torus is made of a Ferroxcube 4C65 material [8] and the yoke from ordinary construction steel. The primary and secondary windings are schematically indicated with a single winding in the construction drawing in order to show their positions. A. Calculation of the Necessary Tuning Range Fig. 5. Magnetic flux lines in the tuning transformer with 13.56-MHz excitation in the primary winding. Due to the high electrical conductivity of the steel yoke, almost no flux generated by the primary coil can penetrate the yoke [4]. The ac flux is effectively decoupled from the steel yoke. dc premagnetization are shown, indicating an almost uniform distribution of the flux lines inside the ferrite core as long as the flux density is below the ferrite saturation (≈0.6 T). This leads to a steady operating point throughout the ferrite with constant permeability, which is adjustable using the control winding. Fig. 5 shows the simulated flux distribution for an RF signal applied to the primary winding. The resulting flux lines show that the alternating-current (ac) flux is concentrated inside the ferrite core and does not at all penetrate the iron yoke. The ac flux in the ferrite is effectively decoupled from the steel yoke. This is due to the high conductivity of the steel material leading to induced eddy currents that prevent the ac flux from entering In order to obtain the tuning range of the inductance of the transformer required to compensate for all possible perturbations, the parameters affecting the resonant frequency have to be considered. The imperfect properties of the components of the setup (tolerance, temperature behavior, and aging) are known from product datasheets. In contrast, the impedance of the antenna system in dependence on objects in the electromagnetic field cannot be easily preestimated. Therefore, we assess a worst case to be expected, i.e., a large massive metallic object in the electromagnetic-field region near the antenna. Finite-element simulations have been performed to evaluate the influence of the metallic plate on the inductance of the whole antenna, for which our tuning unit should compensate. As shown in Fig. 7, the metallic plate is positioned on one side of the antenna, whereas the RFID tag resides on the other side. The antenna is modeled as a simple circular loop antenna with a diameter of 11 cm. Distance d1 between the shielding plate and the antenna is varied for different plate materials, and the variation of the inductance is computed for each parameter set. The results for iron and aluminum plates are shown in Fig. 8. The inductance drops by approximately 35% for the plate positioned near the antenna, as compared with the case 2800 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 60, NO. 8, AUGUST 2011 Fig. 7. Schematic view of the test setup showing the positions of the antenna, the tag, and the shielding plate, respectively. Fig. 9. Inductance of the transductor in dependence on the yoke current. Beginning from a demagnetized ferrite (topmost inductance value), the current was increased to the positive maximum, decreased to the negative maximum, again increased to the positive one, and decreased to zero afterward. (Arrows) Corresponding inductance characteristic. for the lower bound to Tmax = 1.25 · 1.05 · 1.2 = 1.58 −→ +58% (2) for the upper bound. Since a premagnetization can only lower the inductance, the inductor has to be designed to fulfill the requirements at the lower tolerance border Tmin . IV. E XPERIMENTS A. Stationary Behavior Fig. 8. Simulation results showing the variation of the antenna inductance for a simple circular loop antenna with a diameter of 11 cm, depending on the distance of the aluminum or the iron plate to the RFID antenna. The inductance value is referred to the value when the plate is absent. without any conducting object near the antenna. This yields a change in the resonance frequency and a reduced antenna voltage. For the highest inductance variation (∆L = −35%) and a quality factor Q = 30 for the series resonant circuit, the signal voltage is reduced to about 10% of its initial value, thus drastically reducing the maximum distance between the reader and the tag. Even a small inductance variation of ∆L = −3.5% will lead to a significant reduction of the transmitting voltage (−30%) and the achievable reading distance, respectively. The required tuning range for the transductor device is computed to allow for compensating inductance variations of up to 65%. Additionally, the tolerances of the setup itself have to be considered. From the datasheet of the toroid [9], a variation of the inductance factor AL of ±25% is given. From the ferrite datasheet [8], the variation of the initial permeability µi over a temperature range of −20 ◦ C to +100 ◦ C is specified as −13% to +5%. Taking further into account the tolerance of the other components influencing the resonant frequency (capacitor C, undisturbed antenna inductance L, as well as mechanical tolerances of the windings of the tuning transformer), an additional value of ≈20% has to be added. This yields a total tolerance band for the inductance that has to be compensated by the tuning transformer from Tmin = 0.75 · 0.87 · 0.8 = 0.52 −→ −48% (1) In order to measure the tuning range of the transformer, a prototype has been built according to Fig. 6. For the primary RF coil, a three-turn winding is used. The inductance measurement is performed by means of a conventional LCR meter at an operating frequency of 10 kHz. This low measurement frequency is applicable since the inductance of the ferrite is almost constant up to frequencies of ≈40 MHz (see Fig. 1; µ′S in [8]). The dc excitation of the core (i.e., yoke coil current) is controlled by means of a function generator in combination with a power amplifier. Before applying the dc excitation, the ferrite core is demagnetized by an ac flux with a decreasing amplitude. The transfer function of the inductance plotted in Fig. 9 is measured starting from the demagnetized state (the topmost point in the figure). Then, the yoke current is repeatedly changed to the maximum and minimum allowed values to measure the possible tuning range for the inductance, as well as its repeatability. The current limit is given by the maximum temperature increase due to the power loss in the control winding. The realized transductor shows a tuning range of approximately ∆LLCR = 800 nH. This corresponds to a decrease in the initial inductance by ≈40% (from LLCR = 1.92 µH down to 1.14 µH). A larger tuning range could be achieved by using an uncoated core instead of the coated core in the setup presented here. This would reduce the magnetic resistance by lowering the air gap between the steel yoke and the ferrite torus enabling an increased dc magnetization inside the ferrite for the same yoke current. However, for the experiments presented here, the tuning range is adequate, and no modification was required. WEGLEITER et al.: AUTOMATIC ANTENNA TUNING UNIT TO IMPROVE RFID SYSTEM PERFORMANCE 2801 Fig. 10. Equivalent circuit of the tuning transformer. In contrast to a standard transformer equivalent circuit, the main and stray inductances can be tuned by means of the yoke coil current. The values of the secondary side are transformed to the primary one with the transformation factor (N1 /N2 )2 and are therefore marked with an apostrophe. In addition to the inductance change, the performance of the transductor device in the transformer mode was evaluated. The coupling factor between the primary winding and a secondary three-turn winding was determined as c = 60%. With the equivalent circuit of a transformer (see Fig. 10) and the assumption of negligible losses, the circuit can be simplified to a pure inductive divider. The chosen measurement frequency is high enough to avoid losses in the steel yoke (see Fig. 5); the frequency is low enough to avoid hysteresis losses or eddy-current losses in the ferrite. Aside from that, the measured Q factor of ≈25 indicated that our system is almost loss free, too. Hence, the inductance measured by the LCR meter LLCR corresponds to Lσ1 + Lm . Using the inductances in the simplified equivalent circuit, the coupling factor can be expressed as c= Lm Lm = Lσ1 + Lm LLCR (3) from which Lm and Lσ1 can be calculated. Exploiting the symmetry of the transductor device, the secondary stray inductance L′σ2 is equal to the primary stray inductance Lσ1 . Transforming the antenna inductance L to the primary side yields  2 N1 ′ L =L· . (4) N2 Thus, the total inductance as seen by the resonant circuit can be computed from Ltotal = LLCR · L′ + LLCR · (1 − c2 ) . L′ + LLCR (5) For a first experiment, the realized tuning transformer is used in a stationary reader setup. The rectangular antenna measures 30 cm × 80 cm, corresponding to a free-field inductance of L = 2.2 µH. Starting in the midrange of the tunable inductance region (LLCR = 1.5 µH) and the undisturbed antenna (L = 2.2 µH), Ltotal becomes 1.28 µH. A disturbance of the antenna field, as described in Section III-A, decreases the antenna inductance by 35% to L = 1.43 µH, leading to a detuned resonant circuit with Ltotal = 1.22 µH. This relates to a shift of the resonant frequency by   Ltotal −1 ∆f0 = f0 · Ltotal,shifted   1.28 µH = 13.56 MHz · −1 1.22 µH = 329 kHz (6) Fig. 11. Experimental measurement setup (the reader antenna attached to the secondary RF winding of the DUT is not shown, cf., Fig. 1). The generated RF voltage is applied to the primary winding of the DUT by the RF power amplifier. A high-speed oscilloscope records the RF coil voltage and current, as well as the temperature of the DUT. An additional waveform generator is used to apply a premagnetization current to the control winding, which is measured as well. A personal computer controls the system and collects the measured data. leading to a reduction of the antenna voltage by approximately 40%. By adjusting the tuning transformer to LLCR = 1.58 µH, the former value of Ltotal = 1.28 µH can be reestablished, and no reduction of gain occurs. B. Dynamic Behavior To characterize the dynamic behavior of the tuning transformer, a different setup is used [10]. The magnetic properties of the tuning transformer are determined by measuring the amplitudes of the current and the voltage of the RF signal using the setup shown in Fig. 11 (the antenna itself is not shown, cf., Fig. 1). By a mathematical filtering process, the amplitude and the phase of the fundamental harmonics of voltage and current signals are extracted. This gives the complex impedance of the device. In comparison with standard impedance analyzers (LCR meters), which deliver impedance values at rates of up to 10 Hz, our setup can deliver measurement values with 40 kHz. A programmable RF arbitrary waveform generator is used to generate the 13.56-MHz sine-wave signal. An RF power amplifier (100 W; 1-GHz bandwidth) feeds the signal to the device under test (DUT). The voltage is measured at the coil, whereas the current is determined by a current probe. Both signals are recorded using a high-speed oscilloscope (1-GHz analog bandwidth). Additionally, the signal of a temperature sensor, which is mounted onto the ferrite core, is recorded. The impedance values are determined by analyzing the fundamental harmonic of the current and the voltage, and the corresponding phase angle. From the resulting impedance Z, we calculate inductance L and permeability µ; from the voltage, we find the magnetic ac-flux density B inside the material. In principle, this could be done for every single cycle of the RF signal by means of the discrete Fourier transformation. To reach an adequate signal-to-noise ratio, we applied digital filter algorithms and thus reduced the effective measurement rate to 40 kHz, which relates to a temporal resolution of 25 µs. 2802 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 60, NO. 8, AUGUST 2011 The standard stationary reader consumed operational power of about 20 W. Our tuning device additionally required 2 W to almost perfectly compensate for the influence of the applied metallic shield, thus restoring the RFID READ range to its initial undisturbed value. V. C ONCLUSION AND O UTLOOK Fig. 12. Behavior of the tuning transformer when the dc premagnetization is varied over time. (Dots) Filtered measurement values. Fig. 12 shows the measurement results for the complex impedance (absolute value |Z| and phase angle ∠Z) when the dc premagnetization is lowered from an initial yoke current of 103 to 52 mA over a period of 0.1 s. As shown in Fig. 12, the impedance and, therefore, the inductance of the transformer instantly react to a change of the yoke current in a nonlinear and transient manner. The loss factor of the inductance decreases while the yoke current is changed, and it then accommodates to its final steady-state value depending on the dc premagnetization due to the yoke current. The impedance reaches its final value approximately 3 s after the change of the yoke current. In this paper, simulation results have been presented that approve the necessity to retune the resonant circuit of the reader antenna in a passive RFID system, particularly when the reader is moved and the environment of the reader antenna therefore substantially changes. Moreover, retuning the antenna circuit has offered the advantage of automatically compensating for component tolerances and aging effects. For tuning the resonance frequency of the system, we have proposed an electrically tunable inductance device (transductor) whose small-signal inductance can be set by changing the dc magnetization of the ferrite core. Using such a device, experiments have been conducted that demonstrate the capability of the transductor to retune the resonance frequency of the antenna over a wide range. Even for the worst case scenario, the inductance could be successfully tuned, and the system almost reached the undisturbed reading range. Future work will examine in detail the nonlinear relationship between the dc magnetization and the effective inductance, i.e., the incremental and reversible permeability of different ferrite material samples. Particularly for the high-speed tuning of the antenna, the transient phenomena after dc magnetization variations have to be investigated. A sophisticated prediction model for the effective small-signal inductance has to be set up and used to compensate the transient response of the system by driving the dc magnetization coil with a suitable excitation signal. The resulting optimized control strategy will be used to design a stationary RFID reader system. Further developments will focus on miniaturizing the tuning unit to enhance the performance of handheld RFID readers. R EFERENCES C. Transmitting Range Tests The described tuning transformer and antenna were used to build a complete RFID reader system. For our first tests, we implemented a very simple strategy in a microcontroller; the controller adjusts the control current to reach the highest voltage at the resonant antenna, i.e., a simple maximum voltage search. Starting from the undisturbed setup with a manually tuned antenna, a reading range of 103 cm with an antenna terminal voltage of 258 V is reached. Without retuning the resonant circuit, a metallic shield placed 15 cm behind the antenna (≈1/5 of the antenna diagonal) reduced the reading range to 57 cm and decreased the antenna terminal voltage to 102 V. Activating the microcontroller tuning system increased the reading range up to a value of 85 cm; the antenna voltage reached 232 V. [1] K. Finkenzeller, RFID Handbook, 2nd ed. Hoboken, NJ: Wiley, 2003. [2] R. Nopper, R. Niekrawietz, and L. Reindl, “Wireless readout of passive LC sensors,” IEEE Trans. Instrum. Meas., vol. 59, no. 9, pp. 2450–2457, Sep. 2010. [3] B. Jiang, J. R. Smith, M. Philipose, S. Roy, K. Sundara-Rajan, and A. V. Mamishev, “Energy scavenging for inductively coupled passive RFID systems,” IEEE Trans. Instrum. Meas., vol. 56, no. 1, pp. 118–125, Feb. 2007. [4] G. Steiner, H. Zangl, P. Fulmek, and G. Brasseur, “A tuning transformer for the automatic adjustment of resonant loop antennas in RFID systems,” in Proc. IEEE ICIT, 2004, pp. 912–916. [5] P. Fulmek, P. Haumer, and G. Holler, “Hysteresis modelling of NiZnferrites,” in Proc. 31st ISSE, May 2008, pp. 670–675. [6] National Magnetics Group, Inc., Datasheet-NMG M2 Material, Jun. 2010. [Online]. Available: http://www.magneticsgroup.com/pdf/M2 .pdf [7] Y. Bi and D. C. Jiles, “Finite element modeling of an electrically variable inductor,” IEEE Trans. Magn., vol. 35, no. 5, pp. 3517–3519, Sep. 1999. [8] Ferroxcube-A Yageo Company, Datasheet—4C65 Material Specification, Sep. 2008. [9] Philips Components, Product Specification—Ferrite Ring Cores-TN36/ 23/15, Nov. 1997. [10] P. Fulmek, G. Holler, H. Wegleiter, B. Schweighofer, and P. Haumer, “Method for the measurement of transient magnetic ac properties of soft ferrites,” IEEE Trans. Magn., vol. 46, no. 2, pp. 463–466, Feb. 2010. WEGLEITER et al.: AUTOMATIC ANTENNA TUNING UNIT TO IMPROVE RFID SYSTEM PERFORMANCE 2803 Hannes Wegleiter was born in Meran, Italy, in 1981. He received the Dipl.Ing. and Dr.Techn. degrees from Graz University of Technology, Graz, Austria, in 2004 and 2006, respectively. He is a Research Assistant with the Institute of Electrical Measurement and Measurement Signal Processing, Graz University of Technology, with focus on power electronics. Gert Holler (M’04) was born in Graz, Austria, in 1971. He received the Dipl.Ing. and Dr.Techn. degrees from Graz University of Technology, Graz, Austria, in 1999 and 2004, respectively. He is a Project Senior Scientist with the Institute of Electrical Measurement and Measurement Signal Processing, Graz University of Technology, with focus on application-oriented sensing technology and multiphysical simulation. Bernhard Schweighofer was born in Vorau, Austria, in 1973. He received the Dipl.Ing. and Dr.Techn. degrees from Graz University of Technology, Graz, Austria, in 1998 and 2007, respectively. He is a Research Assistant with the Institute of Electrical Measurement and Measurement Signal Processing, Graz University of Technology. Paul Fulmek was born in Steyr, Austria, in 1964. He received the Dipl.Ing. and Dr.Techn. degrees from Vienna University of Technology, Vienna, Austria, in 1990 and 1997, respectively. He is a Research Assistant with the Applied Electronic Materials Department, Institute of Sensor and Actuator Systems, Vienna University of Technology. Christian Deinhammer was born in Wels, Austria, in 1978. He received the Dipl.Ing. degree from Graz University of Technology, Graz, Austria, in 2008. He is a Research Assistant with the Institute of Electrical Measurement and Measurement Signal Processing, Graz University of Technology.