A 5-GHz injection-locked phase-locked loop

3. M.R. Abdel-Rahman, F.J. González, and G.D. Boreman, Antennacoupled metal-oxide-metal for dual-band detection at 92.5 GHz and 28 THz, IEE Electron Lett 40 (2004), 116 –118. 4. M. Abdel-Rahman, F.J. González, G. Zummo, C. Middleton, and G.D. Boreman, Antenna-coupled MOM diodes for dual-band detection in MMW and LWIR, SPIE Proc 5410 (2004). 5. Zeland Software Inc., IE3D simulator, 2004. © 2005 Wiley Periodicals, Inc. A 5-GHz INJECTION-LOCKED PHASELOCKED LOOP Fotis Plessas and Grigorios Kalivas Department of Electrical and Computer Engineering University of Patras Rion 26500, Greece Received 21 December 2004 ABSTRACT: In this paper, we introduce a 5-GHz injection-locked phase-locked loop (ILPLL). A new method is presented for accurate analysis of the phase-noise performance of the proposed system. Furthermore, comparison with other phase-noise-estimation techniques demonstrates that our method provides an accurate characterization of any ILPLL topology. The theoretical and calculation results show an improved performance for phase noise, locking range, and power consumption compared to conventional phase-locked loops (PLLs) and injection-locked oscillators (ILOs). Furthermore, we demonstrate the pulling behavior of the injected oscillator and examine the obtained results. To verify the presented analysis, a 5-GHz prototype has been implemented, which achieves ⫺119-dBc/Hz at 100-KHz frequency offset, producing ⫹4.5 dBm of output power and consuming 9 mA at 3 V. © 2005 Wiley Periodicals, Inc. Microwave Opt Technol Lett 46: 80 – 84, 2005; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.20907 Key words: frequency synthesizer; injection-locked oscillator; injectionlocked phase-locked loop; locking range; phase noise; transfer function; voltage-controlled oscillator 1. INTRODUCTION A promising technique for realizing low-phase-noise levels and increased locking range is the injection-locked phase-locked loop (ILPLL). Studied by several authors [1–5], this method is a combination of a PLL and an ILO. The voltage-controlled oscillator (VCO) of such a loop is an injected oscillator [6 –9], in contrast to the free-running oscillator of conventional PLLs. A microwave oscillator can be synchronized to an external low-power stable signal, thus providing a low-phase-noise high-power output. Figure 2 Common-source capacitive feedback oscillator The input reference signal is divided into two parts using a power splitter. One part is used to provide the reference signal to the mixer, which acts like a phase detector, whereas the other part is injected into the VCO. The mixer produces a phase-error signal that is processed by the loop filter and tunes the VCO. Huang [1] reported the nonlinear analysis of a MESFET injection locked oscillator at 2.7 GHz, whereas Razavi [2] presented the injection-pulling phenomena of a 1-GHz phase-locked oscillator. Optically injected and phase-locked combined systems have been extensively reviewed by Blanchflower [2] and Ramos [3]. Finally, a 10-GHz injection-locked phase-locked oscillator was analyzed and demonstrated in [5], where the same device is used as both phase detector and oscillator. In this paper, we develop a different approach for ILPLL design at 5 GHz by applying a technique used in optical communications [3, 4]. The proposed system, as shown in Figure 1, is suitable for synthesizer applications at 5 GHz due to its implementation characteristics. Furthermore, we newly address the phase-noise analysis of ILPLLs using the loop linear model and compare the results with previously reported work. The analytical expression for the locking range is also derived and the pulling behavior of the injected oscillator is demonstrated. For the experimental investigation, a prototype was implemented using commercial components. The measurement results illustrate the capability of the proposed analysis to accurately predict the phase-noise performance, demonstrate the main characteristics, and confirm the feasibility of the system. The contents of this paper are as follows. In section 2, we present the design and synchronization of the oscillator, which is one of the most critical parts of the proposed circuit. The phasenoise model and locking-range analysis are then proposed in section 3. Section 4 reports the implementation and experimental results, while the conclusions are presented in section 5. 2. DESIGN AND SYNCHRONIZATION OF THE OSCILLATOR 2.1. The Oscillator A common-source capacitive feedback oscillator, illustrated in Figure 2, is employed by using a GaAs Hetero-Junction FET with a parallel resonant applied at the gate. The equivalent circuit of the tank, which consists of a hyperabrupt GaAs tuning varactor (with ␥ equal to 1.25) and an inductor, is shown in Figure 3. C s (V) is the variable capacitor, C p and L s are the parasitic capacitor and inductor, respectively, and R s (V) is the voltage-dependent resistor. Finally, L is the external inductor. The tuning voltage V can vary from 0 to 7 V. Initially, the unstable condition Figure 1 Injection-locked synthesizer 80 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 1, July 5 2005 R tan k ⫹ Rin ⬍ 0 (1) ␻ inj and ␪ inj are the frequency and phase of the reference, whereas ␻ o and ␪ o are the frequency and phase of the free-running VCO, respectively. Moreover, in a PLL with ␻ inj as a reference, the well-known equation for the PLL becomes ␻ inj ⫽ ␻ o ⫹ ⌬ ␻ pllcos共␪inj ⫺ ␪o ⫹ ␸T 兲, (8) where ␸ T is the phase shift around the loop and Figure 3 Equivalent circuit of the parallel tank ⌬ ␻ pll ⫽ K VCOK PDK LF. If we incorporate the injection-locked oscillator into a PLL, the term ␻ o in Eq. (8) represents ␻ inj , as expressed in Eq. (6). The new phase equation becomes is observed. Then R in becomes less negative until R tan k ⫹ Rin ⫽ 0, (2) X tan k ⫹ Xin ⫽ 0, (3) which are the conditions for stable oscillation. Both transient and harmonic-balance analyses are used to obtain the performance of the oscillator. The tuning range of the oscillator is 310 MHz (4770 to 5080 MHz), thus giving ⫹5 dBm of output power. ␻ inj ⫽ ␻ o ⫹ ⌬ ␻ ilpllsin共␪inj ⫺ ␪o ⫹ ␺兲, 2.2. Synchronization The phase noise of the ILO is given by [1]: 冊 2 ␻O VREF LREFcos2 ␸ ⫹ ␻2 LVCO 2Q VO LILO(␻)⫽ , ␻O VREF 2 2 2 cos ␸ ⫹ ␻ 2Q VO (4a) ␸ ⫽ ␪ inj ⫺ ␪ o, (4b) 冉 冊 where ␸ is the stationary phase difference between the freerunning oscillator and the reference signal, ␻ is the offset carrier frequency, V REF , V O , ␻ O , and Q, are the voltage of the reference signal, the voltage of the free-running signal, the free-running frequency, and the quality factor of the embedding network, respectively. LVCO is the single-sideband spectral density of the phase noise of the free-running oscillator and LREF is the singlesideband spectral density of the phase noise of the reference signal. ␸ is expressed as [7]: sin ␸ ⫽ 2Q ␻O ⫺ ␻REF VO , ␻O VREF (10) where ⌬ ␻ ilpll ⫽ 冑⌬ ␻ pll2 ⫹ ⌬ ␻ ilo2 ⫹ 2⌬ ␻ pll⌬ ␻ ilosin ␸ (11) ⌬␻pll cos ␸ . ⌬␻pll sin ␸ ⫹ ⌬␻ilo (12) and tan ␺ ⫽ ⫺ 冉 (9) Eq. (10) is the equivalent Adler’s equation for the proposed system. An increased locking-range is obtained, which is larger than that of a conventional PLL. The numerical results obtained using Eq. (11) are plotted in Figure 4. The locking range of a conventional PLL using the same VCO and phase detector is 18.8 MHz, whereas the locking range of the ILPLL with ⫺25-dBm injected power is 20.4 and 26.1 MHz for ␸ ⫽ 0° and ␸ ⫽ 90°, respectively. For injected-power levels higher than ⫺15 dBm, the locking range of the ILPLL is more than twice that of a conventional PLL. 3.2. Phase-Noise Analysis Let us consider the loop linear model, as shown in Figure 5, where K VCO and K PD are the VCO and phase-detector gains, respectively, and F(s) is the transfer function of the loop filter. The basic components of the loop (injection-locked oscillator, phase detec- (5) where ␻ REF is the reference frequency. 3. LOCKING RANGE AND PHASE-NOISE ANALYSIS 3.1. Locking Range When the VCO (using the ILO mechanism) is locked onto the reference, we obtain ␻ inj ⫽ ␻ o ⫺ ⌬ ␻ ilosin共␪inj ⫺ ␪o 兲, (6) where ⌬ ␻ ilo ⫽ ␻ O V inj , 2Q V out (7) Figure 4 Locking-range characteristics of the injection-locked PLL MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 1, July 5 2005 81 Figure 7 rad2/Hz), the total spectral density of the phase noise of the loop, LOUT(␻), is given by Figure 5 Noise model tor, and loop filter) are assumed to be perfect, noise-free components, followed by a source that introduces noise ( ␸ ILO , ␸ PFD , and ␸ LPF ). If we apply the phase-locked-loop technique to an injectionlocked oscillator, the total output phase noise of the loop is given by ␸ OUT ⫽ 兵1 ⫺ H共s兲其 ␸ ILO ⫹ 1 K PFD H共s兲 ␸ PFD ⫹ H共s兲 ␸ REF 1 H共s兲 ␸ LPF, ⫹ F共s兲 K PFD (13) where G共s兲 and H共s兲 ⫽ . 1 ⫹ G共s兲 Taking into account that LILO(␻), LPFD(␻), LREF(␻), and LLPF(␻) are the single-sideband spectral densities of the noise terms ␸ ILO , ␸ PFD , ␸ REF , and ␸ LPF , respectively (in units of Figure 6 Output phase noise of the proposed ILPLL and a conventional PLL with the same loop filter, reference, and VCO 82 LOUT(␻)⫽LILO(␻)兩1⫺H(␻)兩2⫹LPFD(␻) ⫹LREF(␻)兩H(␻)兩2⫹LLPF(␻) 1 兩H(␻)兩2 兩KPFD兩2 1 兩H(␻)兩2. 兩KPFDF(␻)兩2 (14) When Eq. (4a) is substituted into Eq. (14), the output phase noise of the loop becomes 冉 冉 冊 冊 ␻O VREF 2 LREFcos2 ␸ 2Q VO 兩1 ⫺ H共␻兲兩2 LOUT(␻)⫽ ␻O VREF 2 2 2 cos ␸ ⫹ ␻ 2Q VO ⫹ K VCO G共s兲 ⫽ K PFD F共s兲 s The implemented ILPLL 冉 ␻2 LVCO 冊 ␻O VREF 2 2 cos ␸ ⫹ ␻2 2Q VO 兩1 ⫺ H共␻兲兩2 ⫹ LREF(␻)兩H(␻)兩2, (15) where the effect of LPFD(␻) and LLPF(␻) is not significant enough and have been neglected. The filter is an active integrator with the following transfer function: Figure 8 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 1, July 5 2005 Output spectrum of the free-running VCO Figure 11 Figure 9 tion Measured spectrum of the free-running oscillator under injec- F共s兲 ⫽ s␶2 ⫹ 1 , ␶ 1 ⫽ R 1C, ␶ 2 ⫽ R 2C. s␶1 (16) Computing the overall output phase noise of the proposed system according to Eq. (15), we obtain the results shown in Figure 6. It should be noted that the ILPLL has a 10-dB lower phase noise level at 10-KHz offset from the carrier, as compared to the conventional PLL. The phase-noise improvement can become more significant for integrated (SiGe or CMOS) VCOs. In this case, even 20-dB reduction of in-band phase noise can be achieved, compared to that of a wider loop-bandwidth conventional PLL. A different way of analyzing subharmonic ILPLLs was proposed by Sturzebecher [10], who applied injection locking to a PLL oscillator. We modify the analysis in [10] to suit the fundamental mode (and not the subharmonic used there) of operation, and apply it in our ILPLL. The numerical results we obtain are similar to the results produced by our analysis. These results support the proposed analysis, which constitute an accurate method for the phase-noise characterization of any ILPLL topology. As shown in the next section, the analysis is also validated by the measured data. Phase-noise measurement of the ILPLL Figure 7 was designed and implemented. For the implementation, a PTFE substrate with ␧ r of 6.15 and thickness of 10 mils was used. Impedance matching is achieved by using distributed microstrip-line elements. The prototype dissipates by 27 mW at 3 V. 4.1. The Oscillator The custom-designed VCO, shown within the dashed-line box in Figure 7, was first implemented and tested. A low-noise GaAS HeteroJunction FET and an hyperabrupt GaAs tuning varactor were used, as described in section 2. The resulting measured K VCO was 47 MHz/V. Figure 8 shows the spectrum of the free-running oscillator at 4959 MHz producing ⫹4.73 dBm of output power. Figure 9 illustrates the output spectrum of an injection-pulled oscillator, when ␻ inj is outside the locking range. Razavi studied the pulling behavior of an injection-locked oscillator and arrived at a similar result in [2]. The injected level is approximately 30-dB below the output power level. ␻ inj is the right-hand spectral line (above ␻0). The peak value is at ␻ inj ⫺ ␻ b , where ␻ b ⫽ 公( ␻ 0 ⫺ ␻ inj ) 2 ⫺ ␻ L2 and ␻ L is the lock range of the injected oscillator. The spectrum contains two more sidebands at ␻ inj ⫺ 2 ␻ b and at ␻ inj ⫺ 3 ␻ b . 4.2. The Overall System The phase-noise performance of the 5-GHz reference source taken directly from a signal generator is shown in Figure 10. The power level of the injected signal is ⫺25.46 dBm in this specific case, whereas power levels down to ⫺33 dBm are adequate for efficient 4. IMPLEMENTATION AND MEASUREMENTS To demonstrate the feasibility of the proposed ILPLL and verify the analysis outlined in the previous section, the system shown in Figure 10 Phase-noise measurement of the reference Figure 12 Output spectrum of the ILPLL MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 1, July 5 2005 83 4. R. Ramos, P. Gallion, D. Erasme, A. Seeds and A. Bordonalli, Optical injection and phase-lock loop combined systems, Optics Lett 19 (1994), 4 – 6. 5. H.-C. Chang and R. York, Enhanced MESFET VCO injection locking bandwidth using low frequency feedback techniques, IEEE Int Microwave Symp Dig, San Francisco, CA, (1996), 1515–1518. 6. R. Adler, A study of locking phenomena in oscillators, Proc IEEE 61 (1973), 1380 –1385. 7. K. Kurokawa, Injection locking of microwave solid-state oscillators, Proc IEEE 61 (1973), 1386 –1410. 8. F. Plessas and G. Kalivas, Locking techniques for RF oscillators at 5– 6-GHz frequency range, IEEE Int Conf Electron Circ Syst, 2003, pp. 986 –989. 9. E. Shumakher and G. Eisenstein, On the noise properties of injectionlocked oscillators, IEEE Trans Microwave Theory Tech 52 (2004), 1523–1537. 10. D. Sturzebecher, X. Zhou, X. Zhang, and A. Daryoush, Optically controlled oscillators for millimeter-wave phased-array antennas, IEEE Trans Microwave Theory Tech 41 (1993), 998 –1004. Figure 13 Measurement data superimposed on the calculation results injection-locking to occur. A 4.5– 6-GHz double-balanced mixer is used as phase detector. The measured K PD was 0.4 V/rad. After the characterization of the phase noise of the reference and the free-running VCO, the overall implemented system was measured. The reference signal and the output of the injected VCO are shaped by the noise-transfer functions and added, as presented in section 3. This results in a total measured output phase noise, as illustrated in Figure 11. As expected, the phase-noise curve follows closely that of the reference for frequency offsets lower than the ILPLL bandwidth. Beyond that point, it reduces further to the level of the VCO noise. Figure 12 shows the measured output spectrum of the proposed system, when the oscillator injection-locked and phase-locked, and is relatively clean, and the phase noise characterizing the freerunning signal has been eliminated. A comparison of calculated (solid line) and measured (dashed line) phase-noise characteristics of the ILPLL is given in Figure 13. Good agreement between the measured and calculated curves validates the proposed phase-noise analysis. 5. CONCLUSION In this paper, we have reported an ILPLL suitable for 5-GHz wireless-communication systems. An analytical model was developed and employed to derive the total output phase-noise and locking-range equations. The measurements of the implemented system show that an injected level approximately 30-dB below the output power level is adequate to establish injection locking. The comparison between theory and experiment shows very good agreement. In operation, the proposed ILPLL achieves a phase noise of ⫺120 dBc/Hz at 1-MHz frequency offset, while consuming 27 mA from a 3-V supply. In contrast to conventional PLLs, the phasenoise performance of the proposed system is significantly better, thus making it suitable for integration in modern LAN receivers. REFERENCES 1. C.-C. Huang and T.-H. Chu, Analysis of MESFET injection-locked oscillators in fundamental mode of operation, IEEE Trans Microwave Theory Tech 42 (1994), 1851–1857. 2. B. Razavi, A study of injection locking and pulling in oscillators, IEEE J Solid-State Circ 39 (2004), 1415–1424. 3. I.D. Blanchflower and A.J. Seeds, Optical control of frequency and phase of GaAs MESFET oscillator, Electron Lett 25 (1989), 359 –360. 84 © 2005 Wiley Periodicals, Inc. InGaP/GaAs HBT POWER AMPLIFIER WITH CMRC STRUCTURE C. K. Poek, B. P. Yan, and E. S. Yang Department of Electrical and Electronic Engineering The University of Hong Kong Hong Kong SAR, P. R. China Received 23 December 2004 ABSTRACT: An InGaP/GaAs heterojunction bipolar transistor (HBT) power amplifier is developed for WCDMA user equipment, specifically, band-1-power class-2 application. The HBT power amplifier demonstrates maximum output power Pout of 29.4 dBm and power-added efficiency (PAE) of 48% at a frequency of 1.95 GHz. When operated according to the WCDMA standard, it achieves Pout of 27 dBm and PAE of 32.4%. The adjacent channel leakage power ratio (ACLR) is ⫺33 dBc. A compact microstrip resonant cell (CMRC) circuit is implemented on the HBT amplifier in order to further improve the PAE, ACLR, and IM3 performances. This results in improvements of 8 dB and 6% for the ACLR and PAE, respectively. © 2005 Wiley Periodicals, Inc. Microwave Opt Technol Lett 46: 84 – 88, 2005; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop. 20908 Key words: compact microstrip resonant cell (CMRC); HBT power amplifier; PAE; ACLR 1. INTRODUCTION Current digital mobile-communication systems require high efficiency, high linearity, and low-cost amplifiers for handset transmitters. GaAs-based heterojunction bipolar transistors (HBTs) are a suitable candidate due to their superior power performances. Compared with rival silicon bipolar and GaAs field-effect transistor (FET) devices, HBTs have several advantages, including high linearity, high transconductance, high power added efficiency (PAE), and low 1/f noise. These advantages have made GaAs HBTs the preferred technology for many military and commercial applications. High-efficiency HBT MMIC with a PAE of 40% and a P out of 28 dBm [1] and a high-linearity HBT amplifier module with an output power of 26.3 dBm, a PAE of 50%, and an adjacent channel leakage power ratio (ACLR) of ⫺35 dBc [2] have been developed for WCDMA mobile products. MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 46, No. 1, July 5 2005